Signal equalizer in a coherent optical receiver

ABSTRACT

A signal equalizer for compensating impairments of an optical signal received through a link of a high speed optical communications network. At least one set of compensation vectors are computed for compensating at least two distinct types of impairments. A frequency domain processor is coupled to receive respective raw multi-bit in-phase (I) and quadrature (Q) sample streams of each received polarization of the optical signal. The frequency domain processor operates to digitally process the multi-bit sample streams, using the compensation vectors, to generate multi-bit estimates of symbols modulated onto each transmitted polarization of the optical signal. The frequency domain processor exhibits respective different responses to each one of the at least two distinct types of impairments.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/747,704, filed Jan. 23, 2013, which is a continuation of U.S. patentapplication Ser. No. 13/160,579, filed Jun. 15, 2011, which is acontinuation of U.S. patent application Ser. No. 11/950,585, filed Dec.5, 2007 which issued to U.S. Pat. No. 8,005,368 on Aug. 23, 2011, theentire contents of said applications are hereby incorporated herein byreference.

TECHNICAL FIELD

The present invention relates to high-speed optical communicationsnetworks, and in particular to a signal equalizer in a coherent opticalreceiver.

BACKGROUND OF THE INVENTION

Optical signals received through conventional optical links aretypically distorted by significant amounts of chromatic dispersion (CD)and polarization dependent impairments such as Polarization ModeDispersion (PMD), polarization angle changes and polarization dependentloss (PDL). Chromatic dispersion (CD) on the order of 30,000 ps/nm, andpolarization rotation transients at rates of 10⁵ Hz are commonlyencountered. Various methods and systems intended to address some ofthese limitations are known in the art.

FIG. 1 schematically illustrates a representative coherent opticalreceiver capable of implementing the methods of Applicant's U.S. Pat.No. 7,555,227 issued Jun. 30, 2009 and entitled “PolarizationCompensation In A Coherent Optical Receiver”; U.S. Pat. No. 7,606,498issued Oct. 20, 2009 and entitled “Carrier Recovery In A CoherentOptical Receiver”; and U.S. Pat. No. 7,636,525 issued Dec. 22, 2009 andentitled “Signal Acquisition In A Coherent Optical Receiver”, thecontent of all of which are hereby incorporated herein by reference.

As may be seen in FIG. 1, an inbound optical signal is received throughan optical link 2, split into orthogonal received polarizations by aPolarization Beam Splitter 4, and then mixed with a Local Oscillator(LO) signal 6 by a conventional 90° optical hybrid 8. The compositeoptical signals emerging from the optical hybrid 8 are supplied torespective photodetectors 10, which generate corresponding analogelectrical signals. The photodetector signals are sampled by respectiveAnalog-to-Digital (A/D) converters 12 to yield raw multi-bit digitalsignals I_(X), Q_(X) and I_(Y), Q_(Y) corresponding to In-phase (I) andQuadrature (Q) components of each of the received polarizations.

Preferably, the raw multi-bit digital signals have resolution of n=5 or6 bits which has been found to provides satisfactory performance at anacceptable cost. In the above-noted U.S. patent applications, the samplerate of the A/D converters 12 is selected to satisfy the Nyquistcriterion for the highest anticipated symbol rate of the receivedoptical signal. Thus, for example, in the case of an optical networklink 2 having a line rate of 10 GBaud, the sample rate of the A/Dconverters 12 will be approximately 20 GHz.

From the ND converter 12 block, the respective n-bit signals I_(X),Q_(X) and I_(Y), Q_(Y) of each received polarization are supplied to arespective dispersion compensator 14, which operates on the raw digitalsignals to at least partially compensate chromatic dispersion of thereceived optical signal. The dispersion compensators 14 may beconfigured to operate as described in Applicant's co-pending U.S. patentapplication Ser. No. 11/550,042 filed Oct. 17, 2006, and summarizedbelow with reference to FIGS. 2 a and 2 b.

As may be seen in FIG. 2 a, each dispersion compensator (CD-COMP) 14 isprovided as a high speed digital signal processor (or, equivalently,either an Application Specific Integrated Circuit, ASIC, or a FieldProgrammable Gate Array, FPGA) which is capable of implementing avariety of processing functions. In the illustrated embodiment, twosubstantially identical CD-COMPs 14 are provided, each of which isconnected to receive and process raw In-phase and Quadrature digitalsignals of a respective received polarization. For simplicity only theX-polarization CD-COMP 14 _(x) is illustrated in FIG. 2 a, it beingunderstood that the Y-polarization CD-COMP 14 _(y) will be substantiallyidentical.

In the embodiment of FIG. 2 a, the CD-COMP 14 generally comprises apipelined series of functional blocks, including a deserializer 24, aFast Fourier Transform (FFT) filter 26, a frequency domain processor(FDP) 28 and an Inverse Fast Fourier Transform (IFFT) filter 30.

The deserializer 24 operates to accumulate successive n-bit words of theIn-phase and Quadrature digital signals I_(X) and Q_(X) from theX-polarization A/D converters 12 _(IX) and 12 _(QX) during apredetermined clock period. The accumulated n-bit words are then latchedinto the FFT 26 as a parallel input vector {r^(I) _(X)+jr^(Q) _(X)}.Preferably, each of the real and imaginary components of the parallelvector {r^(I) _(X)+jr^(Q) _(X)} have the same resolution (n=5 or 6 bits,for example) as the raw digital signals. In general, the width (m), inwords, of the input vector {r^(I) _(X)+jr^(Q) _(X)} is selected to behalf the width (M) of the FFT 26. In some embodiments, the FFT 26 has awidth of M=256 taps, which implies an input vector width of m=128complex values. However, a different FFT width may be selected, asdesired. In practice, the FFT width is selected based on a compromisebetween circuit size and the amount of dispersion compensation desired.

The input vector {r^(I) _(X)+jr^(Q) _(X)} is augmented with a nullvector {0, 0, 0, . . . 0} 32 which provides a zero data fill to theremaining input taps of the FFT 26.

The FFT filter 26 performs a conventional FFT operation to generate anarray {R^(A) _(X)} representing the frequency domain spectrum of theinput vector {r^(I) _(X)+jr^(Q) _(X)}. The FDP 28 can then implement anyof a variety of frequency domain processing functions, as will bedescribed in greater detail below, to yield a modified array {V^(A)_(X)}, which is supplied to the IFFT filter 30.

The IFFT filter 30 performs a conventional Inverse Fast FourierTransform operation to yield time domain data 34, in the form of acomplex valued vector having a width equal to the IFFT 30, which, in theillustrated embodiment is M taps. In the embodiment of FIG. 2 a, theIFFT output data 34 is divided into two blocks {v⁰ _(X)}, and {v¹ _(X)},of which {v¹ _(X)} is delayed by one clock cycle (at 36) and added to{v^(I) _(X)+jv^(Q) _(X)} (at 38) to yield the CD-COMP output 16 in theform of a complex valued vector {v^(I) _(X)+jv^(Q) _(X)} encompassingm(=128) complex values.

In the system of FIGS. 2 a and 2 b, the FDP 28 implements atranspose-and-add function, along with dispersion compensation. Ingeneral, the transpose-and-add function operates to add the FFT outputvector {R^(A) _(X)} to a transposed version of itself { R _(X) ^(A)},with respective different compensation vectors. Implementing thetranspose-and-add operation between the complex FFT and IFFT filters hasthe effect of emulating a pair of parallel real-FFT and IFFT functionsthrough the CD-COMP 14, without requiring the additional circuits neededfor parallel real FFT and IFFT filters. The transpose-and-add functioncan be conveniently implemented in hardware, by providing a pair ofparallel paths between the FFT output and a vector addition block 40.One of these paths may be referred to as a direct path 42, in which thetap-order of the FFT output {R^(A) _(X)} is retained. The other path,which may be referred to as a transpose path 44, includes atransposition block 46 which operates to reverse the tap-order of theFFT output, upstream of the vector addition block 40. In this respect,it will be recognised that the transposition block 46 can be readilyimplemented in hardware, which provides an advantage in that thetransposition step does not incur a significant propagation delaypenalty.

Preferably, the direct and transpose paths 42 and 44 are provided with arespective multiplication block 48, which enables various filterfunctions to be implemented by the FDP 28. For example, in theembodiment of FIG. 2 b, a pair of compensation vectors {C⁰ _(X)} and{C^(T) _(X)} 50 are applied to the direct and transpose paths, 42 and 44respectively. Each of the compensation vectors {C⁰ _(X)} and {C^(T)_(X)} is composed of a respective set of coefficients which arecalculated to apply a desired function, in the frequency-domain, to thedigital signals. For example, {C⁰ _(X)} and {C^(T) _(X)} may becalculated to apply a first-order dispersive function to at leastpartially compensate chromatic dispersion of the optical link. {C⁰ _(X)}and {C^(T) _(X)} may also incorporate a transform of a differentialdelay function, so as to compensate residual sample phase errors in theI and Q digital signals. When both of these functions are implemented bythe compensation vectors {C⁰ _(X)} and {C^(T) _(X)}, the CD-COMP output16 will represent a dispersion-compensated and phase-error correctedversion of the raw I_(X) and Q_(X) digital signals received from the A/Dconverters 12.

Returning to FIG. 1, the dispersion-compensated digital signals 16appearing at the output of the dispersion compensators 14 are thensupplied to a polarization compensator 18 which operates to compensatepolarization effects, and thereby de-convolve transmitted symbols fromthe complex signals 16 output from the dispersion compensators 14. Ifdesired, the polarization compensator 18 may operate as described inApplicant's U.S. Pat. No. 7,555,227 issued Jun. 30, 2009 and U.S. Pat.No. 7,606,498 issued Oct. 20, 2009. The output of the polarizationcompensator 18 is a pair of multi-bit estimates X′(n) and Y′(n), 20 ofthe symbols encoded on each transmitted polarization. The symbolestimates X′(n), Y′(n) appearing at the output of the polarizationcompensator 18 are then supplied to a carrier recovery block 22 for LOfrequency control, symbol detection and data recovery, such as describedin Applicant's U.S. Pat. No. 7,606,498 issued Oct. 20, 2009.

In the above described system, the dispersion compensators 14 operatesacross a large number of successive samples (e.g. 128 samples), whichpermits compensation of relatively severe chromatic dispersion, but at acost of a relatively slow response to changing dispersion. This slowresponse is acceptable, because of the known slow rate of change ofdispersion in real-world optical links. The polarization compensator 18,in contrast, is comparatively very narrow (e.g. on the order of about 5samples), to enable a rapid update frequency, which is necessary totrack observed high-speed polarization transients.

The above-described system provides reliable signal acquisition,compensation of dispersion and polarization effects, carrier recoveryand data recovery even in the presence of moderate-to-severe opticalimpairments. This, in turn, enables the deployment of a coherent opticalreceiver in real-world optical networks, with highly attractive signalreach and line rate characteristics. For example, a receiverimplementing the above methods has demonstrated a signal reach of 1500km at a line rate of 10 Gbaud (i.e. 10⁹ symbols/second). It isnoteworthy that this performance has been measured with real-timecontinuous processing, not just burst data acquisition followed byoff-line processing or simulation. The system described above withreference to FIGS. 1 and 2 is the only coherent optical receiver knownto the applicants to have achieved such real-time performance.

With increasing demand for link band-width, it would be desirable toincrease the line rate beyond 10 Gbaud. For example, lines rates of 35GBaud and higher have been proposed. However, as the symbol rate isincreased, the amount of distortion compensation that is required inorder to obtain the same signal reach also increases. For example, therequired amount of dispersion compensation increases proportional to thesquare of the symbol rate, while the required amount of compensation forpolarization effects increases proportional to the symbol rate. Theseincreases in distortion compensation can be met, using the systemdescribed above, but at a cost of increased size and/or complexity ofthe dispersion and polarization compensation blocks.

At the same time, increasing the line rate also necessitates an increasein the sample rate of the A/D converters and downstream digitalcircuits, in order to maintain Nyquist sampling.

It will be appreciated that both increased circuit size and increasedsample rate imply that the power consumption of the receiver mustnecessarily also increase, as will the heat generated by the circuitsduring run-time. This can impose an effective “thermal barrier” toincreasing the line rate, as higher temperatures degrade systemreliability.

Accordingly, methods and techniques that enable reliable operation of acoherent optical receiver at line rates above 10 Gbaud are highlydesirable.

SUMMARY OF THE INVENTION

The present invention addresses the above-noted problems by providing asignal equalizer capable of compensating both dispersion andpolarization, but which is nevertheless agile enough to track high-speedpolarization transients.

Thus, an aspect of the present invention provides a signal equalizer forcompensating impairments of an optical signal received through a link ofa high speed optical communications network. At least one set ofcompensation vectors are computed for compensating at least two distincttypes of impairments. A frequency domain processor is coupled to receiverespective raw multi-bit in-phase (I) and quadrature (Q) sample streamsof each received polarization of the optical signal. The frequencydomain processor operates to digitally process the multi-bit samplestreams, using the compensation vectors, to generate multi-bit estimatesof symbols modulated onto each transmitted polarization of the opticalsignal. The frequency domain processor exhibits respective differentresponses to each one of the at least two distinct types of impairments.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features and advantages of the present invention will becomeapparent from the following detailed description, taken in combinationwith the appended drawings, in which:

FIG. 1 is a block diagram schematically illustrating principal elementsand operations of a coherent optical receiver known from Applicant'sU.S. Pat. Nos. 7,555,227; 7,627,252; 7,532,822; 7,606,498; and7,636,525;

FIGS. 2 a and 2 b are a block diagram schematically illustratingprincipal elements and operations of the dispersion compensation blockof FIG. 1, known from Applicant's U.S. Pat. No. 7,894,728;

FIG. 3 is a block diagram schematically illustrating principal elementsand operations of a coherent optical receiver in accordance with anembodiment of the present invention;

FIG. 4 is a block diagram schematically illustrating principal elementsand operations of the equalizer of FIG. 3;

FIGS. 5 a and 5 b illustrate representative LMS loops for computingpolarization compensation vectors in accordance with a first embodimentof the present invention;

FIG. 6 illustrates a representative LMS loop for computing polarizationcompensation vectors in accordance with a second embodiment of thepresent invention; and

FIGS. 7 a and 7 b illustrate representative frequency domain filtersusable in the LMS loop of FIG. 6.

It will be noted that throughout the appended drawings, like featuresare identified by like reference numerals.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention provides an agile signal equalizer forcompensating dispersion and polarization impairments in a coherentoptical receiver of a high speed optical network. Embodiments of thepresent invention are described below, by way of example only, withreference to FIGS. 3-7.

FIG. 3 illustrates principle elements of a coherent optical receiverwhich incorporates an agile signal equalizer 52 in accordance with thepresent invention. As may be seen in FIG. 3, the agile signal equalizer52 combines the functionality of the dispersion compensation andpolarization compensation blocks 14 and 18 of the system of FIG. 1.Thus, the agile signal equalizer 52 is capable of correcting timingerrors between I and Q sample streams of each received polarization,compensating moderate to severe chromatic dispersion, and compensatingpolarization effects to thereby de-convolve symbols modulated onto eachof the transmitted polarizations from the received signals.

As described in Applicant's U.S. Pat. No. 7,555,227 issued Jun. 30,2009, separating the dispersion and polarization compensation blocks, inthe manner described above in respect of FIGS. 1 and 2, has theadvantage of enabling different compensation response times for eachcompensation block. Thus, the dispersion compensation block 14 is verywide to enable compensation of moderate to severe dispersion, and theassociated slow response (for recalculating the compensationcoefficients C_(X) and C_(Y)) is acceptable because dispersion istypically a slowly changing phenomenon. Conversely, the polarizationcompensation block 18 is, by comparison, very narrow, to facilitate arapid response to polarization rotation transients. Combining bothdispersion and polarization compensation into a common equalizer wouldbe beneficial because it reduces the total number of gates required bythe compensation circuitry, thereby reducing power consumption andassociated heat dissipation problems. However, these potentialadvantages come at a cost of reductions in either or both of dispersioncompensation performance and responsiveness to polarization transients.

The present invention overcomes this difficulty by providing an agilesignal equalizer 52 which has sufficient width to enable compensation ofmoderate-to-severe dispersion. A high-speed Least Mean Squares (LMS)update block 54 provides recalculation of compensation coefficients at asufficiently high speed to enable tracking of polarization transients. Arepresentative coherent optical receiver incorporating the signalequalizer is described below with reference to FIG. 3. A representativeembodiment of the signal equalizer 52 is illustrated in FIG. 4.Representative embodiments of the LMS update block 54 are describedbelow with reference to FIGS. 5-7.

As may be seen in FIG. 3, a coherent optical receiver incorporating thesignal equalizer 52 of the present invention generally comprises aPolarization Beam Splitter 4; 90° optical hybrid 8; photodetectors 10;and A/D converters 12, all of which may operate as described above withreference to FIG. 1. The raw digital sample streams I_(X), Q_(X), andI_(Y), Q_(Y) generated by the A/D converters 12 are then supplied to thesignal equalizer 52. If desired, timing control methods described inApplicant's co-pending U.S. patent application Ser. No. 11/550,042 filedOct. 17, 2006, including the use of elastic stores (not shown in FIG. 3)between the A/D converters 12 and the equalizer 52 may be used to ensureat least coarse phase alignment between samples at the equalizer input.

In general, the equalizer 52 operates to compensate chromatic dispersionand polarization rotation impairments. Consequently, the compensatedsignals 20 output from the equalizer 52 represent multi-bit estimatesX′(n) and Y′(n) of the symbols encoded on each transmitted polarizationof the received optical signal. The symbol estimates 20 X′(n), Y′(n),are supplied to a carrier recovery block 22 for LO frequency control,symbol detection and data recovery, such as described in Applicant'sU.S. Pat. No. 7,606,498 issued Oct. 20, 2009.

In the embodiment of FIG. 4, the equalizer 52 generally follows theconstruction of the dispersion compensators 14 described above withreference to FIGS. 1 and 2. Thus, the raw digital sample streams I_(X),Q_(X), and I_(Y), Q_(Y) generated by the A/D converters 12 aredeserialized (at 24) and the resulting m-word input vectors {r^(I)_(X)+jr^(Q) _(X)} and {r^(I) _(Y)+jr^(Q) _(Y)} latched into therespective X- and Y-polarization FFT blocks 26. The arrays {R^(A) _(X)}and {R^(A) _(Y)} output by the FFT blocks 26 are then supplied to aFrequency Domain Processor (FDP) 56, as will be described below.

The modified arrays {V^(A) _(X)} and {V^(A) _(Y)} output by the FDP 56are supplied to respective IFFT blocks 30, and the resulting time domaindata 34 processed using respective overlap-and-add as described abovewith reference to FIG. 2 a, to yield the equalizer output 20 in the formof complex valued vectors {v^(I) _(X)+jv^(Q) _(X)} and {v^(I)_(Y)+jv^(Q) _(Y)}, each of which encompasses m complex valued estimatesX′(n) and Y′(n) of the transmitted symbols.

In the embodiment of FIG. 4, the FDP 56 comprises a respectivetranspose-and-add functional block 58 for each polarization, and across-compensation block. The transpose-and-add block 58 operates ingenerally the same manner as described above with reference to FIG. 2 b.Thus, the X-polarization transpose-and-add block 58 x operates to addthe FFT output array {R^(A) _(X)} to a transposed version of itself { R_(X) ^(A)}, with respective different compensation vectors {C⁰ _(X)} and{C^(T) _(X)}, to yield intermediate array {T^(A) _(X)}. As describedabove, compensation vectors {C⁰ _(X)} and {C^(T) _(X)} can be computedto at least partially compensate chromatic dispersion of the opticallink and/or to compensate residual sample phase errors in the rawdigital signals generated by the A/D converters 12. Of course, theY-polarization transpose-and-add block 58 x will operate in an exactlyanalogous manner.

The cross-compensation block 60 applies X-polarization vectors H_(XX),H_(XY) to the X-polarization intermediate array {T^(A) _(X)}, andY-polarization vectors H_(YY), H_(YX) to the Y-polarization intermediatearray {T^(A) _(Y)}. The multiplication results are then added togetherto generate modified vectors {V^(A) _(X)} and {V^(A) _(Y)}, as may beseen in FIG. 4. The X- and Y-polarization vectors H_(XX), H_(XY), H_(YY)and H_(YX) are preferably computed using a transform of the totaldistortion at the output of the equalizer 52, as will be described ingreater detail below. At a minimum, the X- and Y-polarization vectorsH_(XX), H_(XY), H_(YY) and H_(YX) impose a phase rotation whichcompensates polarization impairments of the optical signal, and sode-convolve the transmitted symbols from the raw digital sample streamsI_(X), Q_(X), and I_(Y), Q_(Y) generated by the A/D converters 12. Thoseof ordinary skill in the art will recognise that the illustratedcross-compensation block 60 implements an inverse-Jones matrix transferfunction, which compensates the polarization effects. In thisformulation, the vectors H_(XX), H_(XY), H_(YY) and H_(YX) are providedas the coefficients of the inverse-Jones matrix. The width of theinverse-Jones matrix is equal to that of the intermediate arrays {T^(A)_(X)} and {T^(A) _(Y)}, and so is based on the expected maximumdispersion of the received optical signal to be compensated by theequalizer 52.

Preferably, the X- and Y-polarization vectors H_(XX), H_(XY), H_(YY) andH_(YX) are computed at sufficient speed to enable tracking, and thuscompensation, of high-speed polarization rotation transients. This maybe accomplished using the Least Mean Squares (LMS) update loopillustrated in FIG. 4, and described in greater detail below withreference to FIGS. 5 and 6.

FIG. 5 a shows an LMS update loop, according to one embodiment of theinvention, for calculating polarization vectors H_(XX) and H_(YX). Adirectly analogous LMS loop for calculating the polarization vectorsH_(XY) and H_(YY) is shown in FIG. 5 b. In the embodiment of FIGS. 5 aand 5 b, the carrier recovery block 22 operates as described inApplicant's U.S. Pat. No. 7,606,498 issued Oct. 20, 2009. Thus, thecarrier recovery block 22 is divided into two parallel processing paths60 (only the X-polarization path 60 x is shown in FIG. 5 a, and theY-polarization path 60 y is shown in FIG. 5 b),each of which includes adecision circuit 62 and a carrier recovery loop comprising a carrierphase detector 64 and a phase rotator 66. In general, the phase rotators66 use a carrier phase estimate generated by the respective carrierphase detector 64 to compute and apply a phase rotation κ(n) to thesymbol estimates X′(n) and Y′(n) received from the signal equalizer 52.The decision circuits 62 use the phase-rotated symbol estimatesX′(n)e^(−jk(n)) and Y′(n)e^(−jk(n)) to generate recovered symbol valuesX(n) and Y(n), and the phase detectors 64 operate to detect respectivephase errors ΔΦ between the rotated symbol estimates X′(n)e^(−jk(n)) andY′(n)e^(−jk(n)) and the corresponding recovered symbol values X(n) andY(n).

Referring to FIG. 5 a, the H_(XX) LMS update loop receives the phaseerror ΔΦ_(X)(n) of each successive symbol estimate X′(n), which iscalculated by the phase detector 64 as described in Applicant's U.S.Pat. No. 7,606,498 issued Oct. 20, 2009. In addition, the rotated symbolestimate X′(n)e^(−jk(n)) and its corresponding decision value X(n) arealso received from the carrier recovery block 22, and compared (at 68)to obtain a complex symbol error value e_(x), which is indicative ofresidual distortion of the symbol estimate X′(n). In some embodiments itis desirable to format the optical signal into data bursts comprising aplurality of data symbols separated by a SYNC burst having a knownsymbol sequence. In such cases, a selector can be used to supply aselected one of the decision values X(n) and the known SYNC symbols tothe comparator 68. With this arrangement, the selector can be controlledto supply the known SYNC symbol sequence to the comparator during eachSYNC burst, so that the error value e_(x) is computed using the knownSYNC symbols rather than the (possibly erroneous) decision values X(n).

In order minimize calculation complexity through the LMS update loop,the resolution of the complex symbol error e_(x) is preferably lowerthan that of the symbol estimate X′(n). For example, in an embodiment inwhich the symbol estimate X′(n) has a resolution of 7 bits for each ofthe real and imaginary parts (denoted herein as “7+7 bits”), the complexsymbol error e_(x) may have a resolution of, for example, 3+3 bits. Itwill be noted, however, that the present invention is not limited tothese resolution values.

The phase error ΔΦ_(X)(n) is processed, for example using aLook-up-Table (LUT) 70, to generate a corresponding complex value Φ_(X)having a unit amplitude and the same phase as ΔΦ_(X)(n), with a desiredresolution (e.g. 3+3 bits) matching that of the symbol error e_(x). Thisallows the phase error Φ_(X) and symbol error e_(x) to be multipliedtogether (at 72) to obtain a complex vector d_(X) indicative of thetotal residual distortion of the symbol estimate X′(n).

Applicant's U.S. Pat. No. 7,635,525 issued Dec. 22, 2009 describesmethods and systems for signal acquisition in a coherent opticalreceiver. As described in U.S. Pat. No. 7,635,525, during a start-upoperation of the receiver (or during recovery from a “loss-of frame”condition), LO frequency control, clock recovery, dispersioncompensation and polarization compensation loops implement variousmethods to acquire signal, and stabilize to steady-state operation.During this “acquisitions period”, the rotated symbol estimatesX′(n)e^(−jk(n)) and their corresponding decision values X(n) areprobably erroneous. Accordingly, in the embodiment illustrated in FIGS.5 a and 5 b, a window select line may be used to zero out those valuesof the distortion vector d_(X) which are computed from non-sync symbols.Values of the distortion vector d_(X). which are computed from the knownSYNC symbols are likely to be valid, even during signal acquisition, andthus are left unchanged.

In the illustrated embodiments, values of the distortion vector d_(X)are generated at the symbol timing. In the case of Nyquist sampling,this is half the sample rate of the raw digital sample streams I_(X),Q_(X), and I_(Y), Q_(Y) generated by the A/D converters 12, and it istherefore necessary to adjust the timing of the error values d_(x) tomatch the sample timing. In the case of T/2 sampling (that is, thesample period is one/half the symbol period T, which satisfies theNyquist criterion), retiming of the error values d_(x) can beaccomplished by inserting one zero between each successive error value.If desired, Interpolation or other filtering can be performed upon theretimed stream of error values to enhance the loop stability andperformance.

The resulting T/2 sampled symbol distortion vector is then input to aFast Fourier Transform (FFT) block 74, which calculates the frequencydomain spectrum of the symbol distortion vector d_(x).

Preferably, the width of the FFT block 74 corresponds with that of theintermediate array {T^(A) _(X)}. With this arrangement, each value ofthe intermediate array {T^(A) _(X)} can be truncated at 76 to match theresolution of the FFT block output (e.g. 3+3 bits), and then a conjugateof the truncated array multiplied with the FFT output array (at 78), tocompute a low-resolution correlation between {T^(A) _(X)} and the FFToutput. This correlation vector is then scaled (at 80) to obtain anupdate vector {u_(xx)}, which is accumulated (at 82) to obtain a vectorrepresentation of the total distortion of the intermediate array {T^(A)_(X)}. Truncating the total distortion vector, for example by taking the7+7 most significant bits, yields the cross-compensation vector H_(XX).

As noted above, directly analogous methods can be used to compute eachof the other cross-compensation vectors H_(XY), H_(YY) and H_(YX), whichare therefore not described herein in detail.

In embodiments in which the compensation vectors {C⁰ _(X)}, {C^(T)_(X)}, {C⁰ _(Y)} and {C^(T) _(Y)} are computed to compensate onlyresidual sample phase errors in the raw digital sample streams I_(X),Q_(X), and I_(Y), Q_(Y), the symbol error e_(X) will containsubstantially all of the dispersion of the received optical signal 2. Inthis case, the dispersion will propagate through the LMS update loop(s)and the resulting cross compensation vectors H_(XX), H_(XY), H_(YY) andH_(YX) will provide at least partial compensation of the dispersion, inaddition to applying a phase rotation to de-convolve the symbolsmodulated onto each polarization of the transmitted optical signal, fromthe raw digital sample streams I_(X), Q_(X), and I_(Y), Q_(Y).

In embodiments in which the compensation vectors {C⁰ _(X)}, {C^(T)_(X)},{C⁰ _(Y)} and {C^(T) _(Y)} are computed to compensate bothresidual sample phase errors and chromatic dispersion, the symbol errore_(x) will contain only a residual portion of the dispersion. In theseembodiments, the cross-compensation vectors H_(XX), H_(XY), H_(YY) andH_(YX) will provide little or no additional dispersion compensation, butwill still apply the needed phase rotation to de-convolve the symbolsmodulated onto the transmitted polarizations.

A limitation of the embodiment of FIG. 5 is that noise tends to increaseas the speed of the tracking of polarization rotation transientsincreases, e.g. to 50 kHz. It would be preferable to provide low noise,accurate, compensation, while at the same time enabling close trackingof polarization rotation transients of 50 kHz or more. FIG. 6illustrates a modification of the LMS update loop of FIG. 5, in whichthis issue is addressed.

In the embodiment of FIG. 6, the H_(ZZ) LMS update loop of FIG. 5 a ismodified by the addition of a “supercharger” block 84, which is insertedinto the LMS loop between the scaling function 80 and the accumulator82. In this embodiment, it is assumed that the compensation vectors {C⁰_(X)}, {C^(T) _(X)}, {C⁰ _(Y)} and {C^(T) _(Y)} are computed tocompensate at least the majority of the chromatic dispersion, asdescribed above. In this case, the inventors have observed that as thepolarization rotation rate tend towards zero, the intermediate arrays{T^(A) _(X)} and {T^(A) _(Y)} become highly de-correlated with theoutput of the respective LMS loop FFTs 74, and the resulting updatevectors have very low magnitudes. Conversely, as the polarizationrotation rate increases, the intermediate arrays {T^(A) _(X)} and {T^(A)_(Y)} become significantly correlated with their respective FFT outputs,and this is reflected in an increasing magnitude of the update vectors.

The inventors have further observed that under these conditions the timeduration of the majority of a time domain version of the update vector{u_(xx)} is relatively short. This limited time duration occurs becauseof the limited memory inherent in optical polarization effects. The longmemory effects of chromatic dispersion have already been substantiallycompensated, as noted above. Any residual dispersion or other longmemory effects generally only need slow tracking.

The supercharger block 84 exploits these observations by implementing anarrangement in which: 1) portions of the update vector {u_(xx)} that lieoutside the time duration of a polarization effect are suppressed; 2)fully detailed updates are allowed to slowly accumulate, enabling theslow tracking of long memory effects such as chromatic dispersion andline filtering; and 3) the magnitude of the enhanced update vector{u′_(xx)} supplied to the accumulator 82 is scaled in proportion to thepolarization rotation rate.

The suppression of portions of the update vector {u_(xx)} lying outsidethe time duration of a polarization effect reduces the noisecontribution from those portions, and so allows a higher LMS trackingspeed without excessive added noise. However, since this suppression isincomplete, fully detailed updates are allowed to slowly accumulate,thereby enabling accurate tracking of slowly-changing impairments suchas chromatic dispersion and line filtering. Indeed, rather thansuppressing, the illustrated embodiment actually enhances the magnitudeof the relevant time domain portions of the update vector. Finally,scaling the magnitude of the update vector {u_(xx)} in proportion to thepolarization rotation rate effectively increases the update step size ofthe important aspects of the update vectors during high speedtransients, substantially without affecting the ability of the LMSupdate loop to provide accurate compensation (via a small update stepsize) during periods of low-speed polarization rotation.

As may be appreciated, there are various ways in which the Superchargerfunction may be implemented. In the embodiment of FIG. 6, thesupercharger 84 is implemented as a frequency-domain digital filter 86which receives the update vector {u_(xx)} and a summation block 88 foradding the filter output vector {s_(xx)} to the update vector {u_(xx)}to yield the enhanced update vector {u′_(xx)}.

If desired, a threshold block 90 can be inserted at the output of thedigital filter 86, as shown in dashed line in FIG. 6. The thresholdblock 90 can implement any of a variety of suitable linear and/ornonlinear functions to improve loop performance. A low gate-countembodiment is to implement a zeroing function, in which the “raw” filteroutput {s_(xx)} from the digital filter 86 is multiplied by zerowhenever the magnitude of {s_(xx)} is less than a predeterminedthreshold. This can be done individually for each term of {s_(xx)}, orby making one decision for the whole vector based upon a vectormagnitude metric, such as peak absolute value or sum of the squaredmagnitude of each of the vector terms.

As may be appreciated, the frequency domain filter 86 may be implementedin various ways. FIG. 7 a illustrates a low-gate-count embodiment inwhich the frequency domain filter 86 is implemented as a cascade ofsummation blocks. Thus, for example, consider an embodiment in which theupdate vector {u_(xx)} has a width of N=128 taps. These 128 taps can beseparated into K=16 groups of 8 taps each. Within each group, thecomplex values on each tap are summed {at 92}, to yield a correspondinggroup sum B(k). A respective weighted summation value S(k) is thencomputed (at 94) for each group, using the group sum values B(k) of thegroup, and those of the three nearest neighbouring groups. In theembodiment of FIG. 6, for each group k, the weighted summation valueS(k) is computed using the equation

${{S(k)} = {\sum\limits_{i = {k - 3}}^{k + 3}\; {{w(i)}\mspace{11mu} \bullet \mspace{11mu} {B(i)}}}},$

where the weighting factor w(i)=2^(−|k−i|), and modular arithmetic onthe i provides the desirable circular wrap around characteristic.

For example, consider group k=8. The group sum B(k=8) will be the sum ofthe complex values on taps i=64 . . . 71 of the update vector. Theweighted summation value S(k) will be computed as a weighted sum of therespective group sums B(i), i=5 . . . 11. The respective weightingfactor w(i) applied to each group sum B(i) will be w(i)=2⁰=1 for i=k,and then descending by powers of two for each of the three neighbouringgroups. Thus, w(i)=2⁻¹ for i=k±1; w(i)=2⁻² for i=k±2; and w(i)=2⁻³ fori=k±3.

The filter output vector {s_(xx)}, comprising the weighted summationvalue S(k) for each group, is optionally processed by the thresholdblock 90, and then added (at 88) to each of the group tap values of theupdate vector{u_(xx)} to yield the enhanced update vector {u′_(xx)}.Thus, continuing the above example, the weighted summation value S(k=8)will be added back to each of the complex values on taps i=64 . . . 71of the update vector {u_(xx)}.

With this arrangement, the value of S(k) will depend on the degree ofcorrelation between the X-Polarization intermediate array {T^(A) _(X)}and the FFT output vector. When the X-Polarization intermediate array{T^(A) _(X)} and the FFT output vector are highly correlated, S(k) willhave relatively large magnitude (in embodiments in which the thresholdblock 90 is used, S(k) will often be larger than the threshold), and sowill have a strong effect on the enhanced update vector {u′_(xx)},thereby improving the ability of the LMS update loop to track a rapidlychanging polarization angle.

Conversely, when the X-Polarization intermediate array {T^(A) _(X)} andthe FFT output vector are highly uncorrelated (that is, when thepolarization angle of the received optical signal is not significantlychanging), S(k) will have a very low magnitude (in embodiments in whichthe threshold block 90 is used, S(k) will usually be lower than thethreshold, and thus forced to zero), and so will have little or noeffect upon the enhanced update vector {u′_(xx)}, thereby keeping theadded noise to a small level.

FIG. 7 b illustrates an alternative embodiment in which the frequencydomain filter 86 is implemented as an IFFT 96, Time-domain filter (TDF)98 and FFT 100 blocks in sequence. In this case, the IFFT block 96converts the update vector {u_(xx)} to the time-domain, so that the TDF98 can implement a windowing function that suppresses portions of theupdate vector {u_(xx)} lying outside an expected duration of thepolarization effect. The thus “windowed” time-domain update vector isthen converted back into the frequency domain by the FFT block 100, toyield the output vector {s_(xx)}. Various other time-domain filterfunctions may also be implemented by the TDF 98 (either in addition toor instead of the windowing function) as desired.

The above description uses frequency domain LMS. Other adaptive methodscan be used. Zero-forcing is a well known alternative algorithm, whichsuffers from less than optimal noise filtering. Time domain versions ofLMS or other algorithms could be used. This frequency domain version ofLMS has the advantage of a small gate-count and relatively fastconvergence.

The configuration of FIG. 4 can be simplified by omitting themultiplication of the arrays {R^(A) _(X)} and {R^(A) _(Y)} by thecompensation vectors {C⁰ _(X)} and {C⁰ _(Y)}. Mathematical equivalence,to yield identical modified vectors {V^(A) _(X)} and {V^(A) _(Y)}, canbe obtained by dividing the transpose compensation vectors {C^(T) _(X)}and {C^(T) _(Y)} by {C⁰ _(X)} and {C⁰ _(Y)}, respectively, andmultiplying cross compensation vectors H_(XX) and H_(XY) by {C⁰ _(X)},and multiplying H_(YY) and H_(YX) by {C⁰ _(Y)}. In a simpleimplementation, the {C⁰ _(X)} and {C⁰ _(Y)} multiplication blocks in theembodiment of FIG. 4 are omitted. The compensation vectors {C^(T) _(X)}and {C^(T) _(Y)} and cross compensation vectors H_(XX), H_(XY), H_(YY)and H_(YX) are then computed using the techniques described above, whichwill yield the appropriate values.

Other ways may be used for separating the response to slow long memoryeffects from the response to more rapid short memory effects. Patternmatching, transient speed measurement, time moments, error rates,nonlinear equalization, Jones Matrix calculations, and parameterestimations, are examples of methods that may be used, with varyinggate-count requirements. Some of the slower parts of functions could beimplemented in firmware.

Power based scaling or other scaling methods can be used to enhance thespeed of the LMS tracking of the slower frequency components.

The embodiments of the invention described above are intended to beillustrative only. The scope of the invention is therefore intended tobe limited solely by the scope of the appended claims.

We claim: 1-52. (canceled)
 53. A receiver for receiving an inboundoptical signal through a link of an optical communications system, theinbound optical signal having been transmitted by a transmitter as atransmitted optical signal comprising a pair of transmittedpolarizations having a common carrier frequency, the transmittedpolarizations being modulated with transmitted symbols, the receivercomprising: a polarization beam splitter configured to split the inboundoptical signal into a pair of orthogonal received polarizations; asignal equalizer configured to generate estimates of the transmittedsymbols by processing multi-bit digital sample streams of each of thereceived polarizations.